PS-Load Page
Power Supply / Load or 2/4 quadrant voltage current source / meter
Arduino-based Power Supply / Load
The Blog for this project

The Schematics, PCB files, and Simulation models are here

PS-Load 1As an EE, I have designed, used and messed with power supplies for much of my life. My Power-One blog attests to this. As part of this work I have built a number of load circuits from arrays of power resistors to a simple electronic load, to a combined power supply and load circuit. As Keithley and Hewlett Packard and others long ago discovered, a combined power supply and load has some advantages. It combines the functions of a power supply, an electronic load, a voltmeter, and a current meter. As a result it can fully test batteries, solar panels, power supplies, or any power source. When testing batteries, it can charge or discharge using any type of charge cycles. So it is great for battery research and testing. It can either test or simulate a battery. It can test high power LEDs, or other semiconductor devices, and provide V-I curves, similar to a semiconductor curve tracer. An when not providing these advanced functions, can be used as just another lab power supply.

What is the difference between a source-meter and a lab Power Supply? Both can source current and voltage, and measure current and voltage, making them one quadrant devices. The difference is that generally a source-meter is a four quadrant device. I will show that a 2 quadrant device that contains a positive source or load can be a useful configuration.  The number of "quadrants" refers to the polarity of the output voltages and currents, as shown in this diagram.


But only a few power specialized power supplies can sink current. the HP6632A and its siblings are examples of power supplies that can provide a load. Seems like a useful thing to have. Particularly since it only costs a few extra components and a bit more control circuitry over a simple power supply.

To generate a negative power supply, one typically connects a positive supply in reverse, with its + terminal to common (ground) Then it is still a one quadrant device, but a different quadrant.

A source-meter can typically provide both + or - voltages, and can source or sink (+ or -) current while providing either polarity voltage. They are therefore 4 quadrant devices. A source-meter that can only do positive voltage but can source and sink current is a  'two quadrant' device. For testing semiconductor devices, four quadrants is almost a necessity. However, there are advantages to 2 quadrant devices which I discuss below.

A few years back, I built a simple 2 quadrant VI which is described in the Power One page. I have used this to test high power LEDs and batteries. However, this circuit has some limitations. For one, it did not have front panel controls, so it always requires a PC to control it. If the PC crashes or the power goes out during your battery test, you lose data and worse, can over-charge or discharge the battery, causing damage and possibly a fire. Also most power supplies are not designed to drive a battery. If you inadvertently power the supply down, the supply may be damaged by having a high-current voltage on it's output. It is important that any battery teste power down gracefully if the AC is removed.

Keithley source-meters are fine precision devices with multiple current ranges allowing them to source and measure currents from Amps down to microcroamps, and below. They use 5 digit meter circuits, and are generally four quadrant devices. As a result, you pay for these capabilities, typically $4000 for a single channel 2400 unit. If you need to test multiple batteries simultaneously, which is often the case as it takes a  long time to charge or discharge a battery, this gets expensive fast.

For a programmable lab power supply, ADC and DACs are needed, current and voltage measure circuits, a series pass power element (transistor plus heat sink). What I propose here is a combined power supply load that is flexible and accurate enough to perform a range of test functions, and that has the following capabilities:
System and Packaging:
For controls, an optional Front Panel board provides:
A programmable lab supply is one with an external interface such as GPIB or USB. It uses the same basic power and control circuits, but instead of using potentiometers to set voltage and current, it uses DACs, and instead of a digital panel meter to read current and voltage, it uses an ADC. Once a supply is programmable, adding a front-panel consisting of a simple processor, display and controls is straightforward. Ideally a single front panel can control and multiple supplies with a simple menu system.
The other elements of a programmable power supply are a raw DC supply, some low voltage DC supplies for the control circuit, a series pass element and its thermal management (heat sink, fan), control circuitry, a crossover capable of smoothly switching between constant current and voltage control, a reference voltage, and current and voltage measuring circuits. Typically there is a turn-on circuit and status LEDs that indicate whether a channel is on, and in constant current or voltage mode.

To make a 2 quadrant PS-Load, a second series pass element is required. Since only one series pass element is drawing current and therefore dissipating power at a time, it can share the heat sink and fan with the other device. Another DAC to set the negative current is optional. My original 2 quadrant design used a single DAC to set the + and - currents to the same value but opposite polarity with good successes. Some commercial products also use this shortcut.

To make a 4 quadrant device, the raw supply provides both + and - voltages. The advantage of 4 quadrants is that it can provide either + or - voltage under user control. A disadvantage is that it may be easier to fry a DUT by inadvertently applying a negative supply. A subtle but important downside of a 4 quadrant system is that when using it as a load, the current for the load is generally derived from the negative supply, not the ground return. The negative pass element  therefore has the negative raw supply voltage plus the positive load voltage across it. This burns much more power than a normal electronic load, or a 2 quadrant device where the negative pass element is simply grounded. So the circuit tends to be power limited instead of current limited, and therefore cannot sink as much current.

So if you plan to use your PS-Load as a high power load such as a battery or power supply tester, the 2 quadrant version is preferable, and also slightly lower cost.

Output Stage

There are many output stage designs that can be used including bipolar transistors and FETs. Here are the general output stage requirements:
Emitter followers using bipolar transistors can meet most needs except for the low voltage requirement. FETs can achieve < .1 ohm to GND as a load.

Complimentary, common-drain MOSFETs can pull the supply very close to the rails (low drop-out), but have a high output impedance (essentially a current source) until they come close to saturation. This high impedance means that the amplifier loop gain is a strong function of both the load resistance and capacitance. This makes it difficult to stabilize the loop for all kinds of loads when these types of stages are used.

Complimentary Source followers have low output impedance, but require gate drive circuits that exceed the power supplies, or else can have up to 5V loss at high currents. I call the high voltage supplies V++ and V-- boost supplies. These can be built with a simple charge pump consisting of a 555 oscillator and some diodes and capacitors. This type of stage requires that the driver amplifier have a fairly high power supply voltage rating: the raw supply (+/- 25V) plus about 12V extra on each supply is 74V. There are several amplifiers that meet this need. To accommodate even higher supply voltages or unregulated supplies, the supply requirements are higher. Linear has a nice high-voltage opamp, LTC6090 that accommodates up to 140VDC and costs about $3 (q100).

output circuit
Simplified Control and output circuit for single quadrant (+24V) version: The shut-off integrator control circuit is not shown here.

My initial spec of +/- 24V output range can be met with a +/- 25V power supply. These can be met with low-cost 24V switchers adjusted to 25VDC. This allows 0.5V drop for the current shunt and 0.5V for the FET, wiring, and fuse. Figure 0.25V at 5A for the FET, or 50 milliohms.

This type of output stage is quite simple. Both FET's gate pins are tied together, and both source pins are connected. A single resistor from both Gates to both Sources will turn off both FETs. Apply a positive voltage and the + NFET turns on. Apply a negative voltage and the - turns on. Another advantage is to use a simple passive switch between the voltage amplifier stage and the FET gates to turn OFF the output stage.

This type of output works well when one FET is on and when the load changes quantity. But in the case when the load current changes polarity, the output stage takes a few hundred uS for the drive voltage to change the few volts from turning on one FET to turning on the other. In an audio amplifier, this load current polarity change occurs on every cycle of AC, and would cause horrid crossover distortion. In the case of a Voltage / current source, it occurs rarely and the circuit recovers quickly.

Crossover Design: Single integrator

Crossover design refers to the current and voltage control circuitry. It is called a crossover because it must cleanly switch from voltage to current control and vice versa. There are a lot of ways to implement this. Most basic power supplies use two integrators, one for voltage and one for current. Then diode logic is used to control the pass element with current control taking priority over voltage control. The disadvantage of this type of control is that when the loop is regulating in one mode, the other integrator goes to its maximum output. Then when the load or setting is changed and the other integrator takes control, the output voltage or current can overshoot while waiting for the other integrator to slew to the correct value. Overshoot is generally bad in a power supply. This problem only gets worse with a 2 or four quadrant device. A third integrator is required for the load side (quadrant 2) or -current control. 

The approach I took uses a single integrator with active diode logic to select the error voltage from either the voltage, +current, and - current sources. There are three error amplifiers each consisting of a basic op-amp plus a diode. These slew quite fast to control the single integrator and so overshoot is minimized. 

Power-Off and safety

If you analyze the circuit of a typical power supply, you will see that things get complicated if you apply an external power supply (battery) and then turn off the supply. I have toasted a couple of power supplies doing this, and since I want this PS-Load to be able to test batteries and other sources, I designed it from the start to be able to handle this condition gracefully.

To turn off the supply, the output stage must be somehow disabled. with a simple load, simply setting the output to 0.0V may be adequate. But when the load is another voltage source, the output current must be reduced to zero. The complimentary FET output  stage does this nicely, with the single gate resistor nicely turning off both FETs. But it is necessary to somehow isolate the gates from the driving amplifier. I chose a Photomos(TM) FET switch for this function. These $1 devices have the advantage of requiring no power supply pins, and so to turn them off, simply do not drive the LED input. So when the power supply is off, the FET is guaranteed to be off. Perfect.
A relay would also provide this function. The Photomos devices are smaller and never wear out. The typically switch in about 1mS and handle milliamps up to amps, and will switch 100 or more volts.

But when the Photomos is turned off, what about the integrator? The control loop is now broken, so the integrator will slew to one extreme or the other. Then when you turn the switch back on, the output voltage will have a huge glitch. You could reset the integrator to 0V, but if the load is a 12V battery, the turn-on current transient will be huge until the loop recovers. Not good. One solution, used by HP in their excellent HP6632A supplies, is when the output is OFF, to control the integrator to output the same voltage that the output pin is. For example, if the output is a 12V battery, the output of the integrator and driver op-amp is controlled at 12V. Then when the Photomos is turned back on, the FETs have no gate voltage and are initially off. This elegant solution  works very well both in simulation and in reality.

Speaking of Simulation

I generated an LTSpice simulation for this design. Do not underestimate the complexity of a lab power supply. They must remain stable over a wide range of resistive and capacitive loads, and must switch from constant voltage to constant current cleanly and without overshoot. Designing and testing circuits for all of these conditions is work. It is much easier to test them with a simulator and observe the step response. Then when the simulation performs well, build and test a real circuit.

Current Sense: Many possible configurations

High side or low side current sense? High side means that the current sense resistor is in series with the + output of the supply. Low side means it is in series with the ground. High side is required if you have multiple supplies operating from the same raw supply. Low side is simpler since the sense amp doesn't need to deal with common mode voltage changes.

With the hundreds of current sense ICs on the market, this should be easy, no? No. All current sense chips require a minimum voltage of a few volts to operate. With a lab supply where the output voltage can be 0V, they won't work without tricks. And forget about measuring current when the voltage is negative. So they don't work for either high side or low side applications.

With a high side current sense amp, common mode is an issue. Take a typical application:a 0 to 25V power supply with a .1 ohm 5A current sense. 5A is 0.5V across the sense R, and if you want about 1mA accuracy, that is 100uV across the R. If the voltage can be 0 to 25V, and you don't want the current reading to change more than 1mA, that is 25V / 100uV = 250,000 :1 or 108 dB Common mode rejection ratio (CMRR). Yikes! Some instrumentation amplifiers can do this, but the problem is that the amplifiers require high voltage power supplies. If you use a resistive differential amp, the resistor matching is from hell. You can trim the CMRR with a trimpot, but 250:000:1 is tough. Normal 1% resistors are spec'ed at 100ppm drift per degree Celsius. 1/250,000 is 4ppm total drift over the full temperature range. 

I used a compromise: 25ppm 0.1% resistors (available for about $.20) and a nice 20 turn trimpot. The circuit works quite well but is not designed to hold that CMRR across a wide temperature range.

Turn-On/Off circuit

The Turn-On/Off circuit has a few requirements. When PS-Load is OFF, I want the OFF current to be as low as possible, preferably just a few uA. I like the nice simple approach of using complimentary FETS as output stages, with a simple resistor to bias them OFF. This requires some isolation from the voltage amp when in the OFF state. I use a Photomos solid-state relay. If you haven't used these, they are pretty cool for low and medium current switching. They are optically isolated and consist of an LED plus MOSFET switches. They have on-resistances in the 50 down to 0.1 ohm range, can output AC or DC switch voltages up to 100's of volts, and provide thousands of volts of isolation from the LED to the switch. They are available from a handful of manufacturers and with safety ratings for AC line applications. They are like a relay but use less voltage and current to drive: 1.2V at a few milliamps. They are more reliable than relays with no moving parts or contacts to wear out. They are small in a 4 pin SO or DIP package, and cheap, about $1. Like a relay they do not turn on or off instantly, but take about 1 millisecond, similar to a reed relay. Larger armature relays take 10s of milliseconds.

So one of these is used to connect / disconnect the voltage amp to the FET gates.

When the circuit turns ON, the output voltage should slew cleanly from its idle voltage to the voltage setting. For example, imagine a 12V battery connected to the output. When the output is OFF, I want minimum load on the battery. When the supply goes ON, I want minimum transient current to occur. When OFF, the output FET gates are held at the output voltage by the bias resistor. But when ON, the gate voltage is switched to the high voltage amp output voltage. If that voltage was for example 12V, this would turn ON the P-Fet hard for a few milliseconds until the current and voltage loops could take over. Turning a FET on hard between Ground and a 12V battery will toast it due to the extremely high instantaneous current. Hopefully the output fuse would blow before the FET, but nuisance fuse blowing is also bad.

The fix for this problem is to hold the high voltage amp output near the supply output voltage when the supply is OFF. Then when the Photomos is turned ON, the gates and sources of both FETs are at the same voltage, and the FETs are initially OFF. A separate  voltage divider measures the buffer amp voltage and compares it to the Output voltage. This error signal. Vbuf - Vout controls the integrator when the supply is OFF, holding the buffer at the output voltage. A CMOS switch, DG419, switches the integrator input between normal operation and this difference signal. A logic signal called ON from the I2C bus, controls this CMOS switch and also turns on the Photomos.

I finally got a nice digital scope and can now measure power supply transient behavior. When I hit the ON button, with PS-Load set to 5.0V, I saw a nasty +15V transient for about 300uS. Definitely bad power supply behavior. When the ON signal goes ON, two things occur. The integrator is switched from the voltage amp to the to the error amps, and the output stage is connected to the voltage amp. But the DG419 switch for the integrator switches in a few nS, while the Photomos switch takes about 500uS to turn on. So the integrator starts slewing right away, but the loop is broken for 500uS. The fix is to make the integrator control input (DG419) as slow as the Photomos. So I added a 500uS RC delay to the DG419 control signal.


In the inevitable discussion between I2C and SPI, there are many trade offs. I generally choose I2C where speed can be slow, and isolation, and multiple addresses (devices) are required. So for a fast ADC,  use SPI, but for this relatively slow Sigma-Delta ADC, I2C is fine. I need DACs, a multi-input ADC, general purpose IO (GPIO) and an EEPROM to store setting and calibration data. In addition, I might use 3 or 4 identical PS-Load boards in a system with a single processor controlling it all. So one parameter I looked for in selecting the I2C peripherals was the ability to easily select 4 I2C addresses per each device.


A system power supply needs to be isolated from the computer system that is controlling it. Imagine a USB interface to a non-isolated power supply, and the operator accidentally shorts the 24V 3A output to ground. That will cause 3A to flow through the AC ground pin, or worse, the ground pin of the USB cable to the PC. Not good. For this and other reasons, isolation is needed. 

I2C isolation is a bit tricky since at least the Data wire needs to be bi-directional. For multiple masters (rare) the clock would also be bidirectional. Fortunately there are I2C specific isolators that address this trickiness. I use the ISO1541 from TI in a SO8 package for about $4.

Power Supplies

The system can use any raw power supply including low-cost switchers, and conventional transformer / diode / caps.  To generate 24V, I use a low-cost 24V switcher set to +25.5V so the supply can output 24V at full current, including wiring, shunt resistor and FET losses.

In addition to the raw supply, the FETs need "boost" supplies. I discussed these earlier.

In addition to these, the analog circuits need +/- 12V for the analog stuff,  +5V for the I2C stuff, and an isolated +5V for the control processor.
For +/- 12V, I use a low-cost DC-DC from CUI. The 5V is a fairly low current (a few milliamps) so I use a 5V linear regulator from the +12V. For the isolated 5V to drive the processor, I use a small, isolated DC-DC.  Note that any isolated 5V supply can be used. In fact if multiple channels are used, only one isolated 5V is required. and so channels 2 though 4 should not have the 5V DC-DC installed.


After using the Microchip16/18 bit 4 channel Sigma-Delta ADCs (MCP34xx) on previous projects, I chose it for the ADC on this project. 16 bits at 7.5 samples per second, but the cool feature is the ability to measure +/- 256mV with a single +5V supply. Bipolar current and voltage measurements need bipolar measurements. The part only uses an internal reference, and doesn't bring it out as a pin, but it's 15ppm/C drift is quite good. I only use 2 of the 4 channels in this design, but the 2 channel parts don't have I2C addresses, so I use the 4 channel part here.


DAC INL (Integral Non-linearaity) is a critical spec for a programmable voltage source. It is important to get an accurate value when you request it. Offset and gain errors can be calibrated out, but to get accurate values at all settings, low INL is needed. Many system power supplies use expensive14 or 16 bit DACs for this purpose. I need 2 or 3 channels, one for Voltage, one for Positive current, and one for negative current. I need 12 or more bits with good INL, and low cost. Unfortunately most low-cost 12 bit bit DACs have high INL, like 4-12 LSBs. 12 or more LSBs of INL is equivalent to an 8 or 9 bit DAC. I found the Maxim MAX5815 quad 12 bit DAC that has +/-1LSB of INL Max. and is about $5. Yes, it has address select pins.  And they throw in a sweet 10ppm/C reference. I use it for the other circuits that need a stable voltage.

Another way to get better INL is to use the more accurate and higher resolution ADC as feedback, and adjust the DAC setting. With this approach, the initial setting will have the full DAC INL error, but then the ADC corrects it. This causes a somewhat strange settling response as the firmware control loop hunts for the best DAC setting. I'm not sure this is an acceptable behavior.


There is an I2C EEPROM per channel that will ultimately contain the calibration factors for that channel. But implementing proper cal firmware and EEPROM storage is a lot of work, which I have not done yet. For now, I simply hard-code the cal factors in the Arduino firmware. Since I currently only have 2 boards built, this is a reasonable approach.

For each board, there is a table like this:

#ifdef BOARD1

  Set[DVS].range_max = 25.0;  
  Set[DVS].set_max   = 25.0;
  Set[DVS].range_min = 0.0;
  Set[DIP].range_max = 5.0;
  Set[DIP].set_max   = 3.0;
  Set[DIP].range_min = 0.0;
  Set[DIN].range_max = 5.0;
  Set[DIN].set_max   = 5.0;
  Set[DIN].range_min = 0.0;
  // Offset and Gain for all ADCs and DACs
  Set[DVS].gain_cal = 0.939145;
  Set[DVS].offs_cal = -0.022;
  Set[DIP].gain_cal = 0.99475;
  Set[DIP].offs_cal = -0.022;
  Set[DIN].gain_cal = 1.00000;
  Set[DIN].offs_cal = 0.0000;
  Measure[VM].offs_cal = 0.004;
  Measure[VM].gain_cal = 0.99534;

  Measure[IM].offs_cal = -0.011;
  Measure[IM].gain_cal = 0.99161;

  Measure[VM].range_max = 13.00;     // 2Q: /13 only    
  Measure[IM].range_max = 20.0;     // .1 ohm = 10V/A

  Set[DIP].value = 3.000;           // Set initial current values
  Set[DIN].value = 3.000;

To begin calibration, I set the offsets to 0.00 and the gains to 1.00. I start by calibrating the16 bit Voltage ADC: apply 0.0V to the output (a short or low value resistor, power supply OFF) and see what the voltage display reads. Lets say it reads +0.015V. So the voltage measure offset,
Measure[VM].offs_cal is -0.015V.
Then set the output to about +20V and compare the measured (display) and expected (DMM) values. Divide them, and that is what you wold need to scale the measured by to obtain the expected. Set 
Measure[VM].gain_cal to that number. Do the same for the current readings.

Once the measure circuit is working, you can use it to measure and calibrate the voltage DAC and two current DACs. I generally enter the offset number and recompile before calibrating the gain. More compiling, less thinking. The first time you do this will will take a bit of time, but once you get the hang or it, maybe 20 minutes to calibrate everything. Notice that I got lazy and did not cal the negative DAC [DIN] yet.


My goal is to minimize case size. I would like to fit a single channel into a small 2U (3.5") high by 1/4 rack (4.25") wide case. Length should be 10-12". I came across a large batch of free Hammond 1402F enclosures. These are 7x10x3", and fairly attractive. Here is a single channel system in a Hammond case. Note the switching supply beneath the PS-Load board. For an AC inlet I use a fused version, although switching supplies already have AC fuses. If the design uses a transformer, then a fuse is required. Also since I run AC to the front panel, an AC fuse is a good idea.
For a heat sink, I used a 2.6 x 1.5 x 2.4" one cut from a larger heat sink I found on Ebay. It is a bit too small to dissipate 50-75W, so I currently run at a bit lower power. The 24V fan blows air out of the enclosure, which I think is not quite as efficient as blowing cold air directly onto the heat sink. I still have some work to do here. Ideally I would find a larger heat sink. Note that for two or three channels, a single wider heat sink could be used and multiple boards could be mounted side-by-side. Currently the height of the heat sink and the switcher are about the same, so I just mount a couple of short spacers on the switcher to hold the front two mounting holes of the board. It's a bit of a hack, but it works. It requires removing the cover of the switcher to mount the spacers, nuts, and washers. You could also just use long spacers for the board. The switcher fits between the board mounting holes.

Internal view with the original heat sink and no fan board

New Heat Sink
New heat sink with thermistor and fan board installed.

The original heat sink was a bit small and did not cool efficiently. Too much of the cooling air diverted around the heat sink. Also it had no mounting features. I purchased another large heat sink extrusion on Ebay, cut it to size on my trusty table saw, and mounted it to two small angle brackets. It is raised about .25" off the chassis bottom to allow air to flow efficiently past it. This new one works well. Ideally there would be fins above the heat sink as well as below it. Here is an off-the shelf Wakefield 395-1AB that is about the right size should work nicely and costs about $22.

Front panel controls, Encoder, push buttons and LCD. No Green binding post yet.

Fan Control 

A thermally controlled fan is generally good. When the supply is not outputting much power so the heat sink is cool, the fan can be OFF. When it is operating at max power, the fan is ON full. In the middle it wold be nice to have th4e fan ON but quiet since no-one wants to hear a fan that is on unnecessarily or is constantly varying in speed, the sound is annoying and distracting. So  the speed should change only when it needs to. It's a trade-off between optimal cooling and noise. My design is for 3 fan speeds: Off, full on, and medium. This should do the job and minimize fan speed changes.

Most DC fans are 12V, but I have lots of 25V power and little 12V, so use a 24V fan. But the fan controller should also accept 12V power and drive a 12V fan. I might want to have multiple heat sinks or a large one with multiple temp sensors. I might want to have 2 fans. Since the 24V is electrically isolated from the CPU or I2C control, any feedback or controls from the CPU to the fan would need to be isolated.
Fan power isolated from GND and other supplies I considered a few approaches, mostly by having the Arduino control the fan. Finally I decided that boards are cheap and I just needed a variable fan. So I built an analog circuit that can accept 1 through 4 thermistors, drive multiple fans, and has 2 setpoints for 3 fan speeds. Here is the schematic and PCB. By the way, this is my first PC board order from $10 for 5 boards, $8 shipping, 3 weeks delivery.

The maximum temperature circuit is actually a minimum voltage circuit since high temperature on an NTC thermistor causes low voltage. It uses a quad op-amp and 4 schottky diodes to output the minimum of 4 thermistor voltages, representing the hottest of the 4 thermistors. I looked up the two temperature setpoints, 40C and 55C. The desired fan voltages are 0V, 12V and 23V. 12V is pretty quiet, 23V is the maximum that the LM317 regulator can output with a 25V input. These temperature setpoints correspond to voltages when used with a 5V supply and a 10K bias resistor, or
1.50V and 1.25V. Then I built a simple 3 resistor chain to provide these 3 voltages. The temperature and the setpoints are compared via two LM393 comparators, which drive resistors that set the LM317 output voltage.  Whew!

The BOM cost of the board with a single thermistor and one fan output is about $2 plus $1 for the board. For thermistors, I use Lug mounted. These are a bead type thermistor mounted in a copper lug that can be mounted to a surface with a single screw. They are about $1.

Fan Board


The schematic was designed in DipTrace. After considering the usual low-cost CAE options, I decided on DipTrace. Unlike most CAE tools, I like it.  For years I used ExpressPCB. They were the quickest and lowest cost way to get prototypes. But their file format is proprietary, and many offshore vendors now produce boards much cheaper than ExpressPCB, and require Gerber file formats. It was time to update my skills to a real PC design tool.

I attended a Synthesizer conference at Olin College in Needham, MA. I spoke with an EE professor there and asked him about building prototypes for students. He said that he regularly assigned a schematic and PCB project to students to do over a weekend. Students who had never drawn a CAE schematic or laid out a PC board could download and learn DipTrace quickly. When I tried it, I was pleasantly surprised and have been using it for the past 8 months.



The BOM has costs included. The parts on the PS-Load board are about $50, Qty 100, pretty reasonable. In addition, A 75W 24V, 3A switching power supply is $27. The heat sink cost is about $5-10. I buy surplus heat sinks on Ebay and cut them to size. Then a case, 24V fan, AC inlet, AC switch, and banana jacks.

For the front panel board, a PCB, LCD, Atmel processor, crystal, USB connector, a couple of buttons and an encoder. I estimate $20 cost.

Front Panel Board 

I used the front panel board of my Arduino Panel Meter for the processor and controls. In addition to an Arduino Leonardo compatible (ATMega32U4) processor, it has an 8x2 LCD, an encoder knob and positions for push button switches. The panel Meter has isolated data acquisition circuitry which I bypassed, and brought out its I2C circuitry. It was a good match for this application. While the small 8x2 LCD is OK for a single channel unit, it is a bit too small for a multi-channel system, and the digits are fairly small. My plan it to lay out a 16x2 version that can use either a small or large digit LCD.


I wanted to use all Arduino code for this project to make it as accessible as possible. I like that Arduino can do basic math in floating point and still fit nicely in a 32KB memory. The code uses the wire library for I2C, and the LCD library. One problem with Arduino libraries is that they tend to use polled methods as opposed to interrupts. I like to read front panel controls including encoders in a 1mS or so timed interrupt routine. Fortunately the Arduino also uses a 1mS timed interrupt using Timer0. I found a trick of adding a second output compare to Timer0, which creates a second interrupt. You can adjust the compare value so the second interrupt occurs roughly halfway between the Arduino interrupts, thus avoiding conflict. Then you can add your own code to this new interrupt. I used this technique to create an interrupt where I poll and debounce the keys, and to poll the encoder. Keep in mind that this interrupt isn't exactly 1mS, but  is 16Mhz / 64 / 256 = 976.5625 Hz. Arduino timer routines compensate for this error.

Switching Power Supply Common-Mode Noise Issue

With any power supply, either lab or otherwise, common-mode noise is an issue. When you float a power supply, there is always some AC current flow from the power supply ground to the chassis (AC) ground. With a linear supply, this is usually a small amount of 60Hz current due to the inter-winding capacitance between the primary and secondary windings of the power transformer. With a linear supply, the frequency (60Hz and some harmonics) and the capacitance (20-200pF) and the 240VAC input causes about I = V / Xc = 240/(1/2*pi*60Hz * 200pF) or tens of microamps. No problem, and the typical .01uF safety cap to ground shunts out most of this current.  In the case of a high-class power supply or a precision instrument, the AC transformers are usually double or even triple shielded with metal foil between the windings. This shielding reduces the common-mode noise current significantly.

However with a switcher and its high frequency transformer, the frequency is not 60Hz, but the harmonics of the the fast rise-time switching waveforms: rise times of 300V pulses can be about 100ns causing pulses with harmonics of 10MHz or more. The transformer windings are usually smaller, so the inter-winding capacitance is a bit less. Just to meet radiated and conducted EMI, the transformers are often shielded. You will sometimes see copper foil on switching transformers.

Still, I see some pretty ugly looking common-mode switching noise on many switchers.  Manufacturers do not specify common-mode noise, so how do you deal with it?
How do you quantify it? Search for this problem on line and you will find no specific data or techniques. In fact, to meet EMI, Switchers are often tested with a short and heavy wire from their DC common to the chassis ground. This effectively shunts any common mode noise to ground. But if your application requires a floating supply, you are on your own dealing with this issue. Measuring the open-circuit voltage is interesting. You will typically see a few volts of high frequency crud. Why not 100V, since the switcher is switching hundreds of volts? The answer is that switchers do have a capacitor from AC to DC ground, typically between .01 -.05uF. This is a safety rated cap in case the ground of the system is accidentally opened up.

This common mode voltage or current can shows up as noise on audio or other critical analog circuits. It basically causes an unavoidable AC ground loop at high frequencies. If you need a switching supply to float, beware.

To measure the common-mode noise of a power supply, I use a simple current measurement. A 10 ohm resistor has bandwidth out to the GHz range. Wire a 10 ohm 1/4W resistor between the chassis ground pin and the DC common, usually V-. Measure the voltage across this resistor with a 20MHz or 100MHz scope, and you have a good indication of the high frequency common-mode currents flowing through the supply. I did this on several switchers and as expected, most had about 1V p-p of crud across 10 ohms or 100mA of switching currents.  But to my surprise, I found some switchers are quiet, measuring less than 20mV across 10 ohms or just 2mA! What do they know that the other guys don't?

To investigate this, I first measured the capacitance from GND to V-. All supplies measured about .02uF, meaning that the manufacturer typically uses a .022uf capacitor there. So I opened up the bad and two good ones to see what the difference was. The bad ones use a safety-rated, thru-hole, ceramic disc cap from V- to GND. Seems reasonable. But the good ones use an array of 3x2 surface mount capacitors and much shorter and thicker PC traces.  And they mount the capacitors directly between the V- and GND pins of the supply. This approach minimizes the circuit inductance and therefore the high frequency noise. Nice.

Who is good and who is bad? All the CUI (V-Infinity) supplies I measured (n=3) were bad. All of the TDK-Lambda supplies (n=2) were good. I will be using TDK-Lambda switchers from now on when I am concerned about noise.

Photo and schematic of test circuit
PHOTOs of good and bad noise
PHOTOs of good and bad layouts

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Last Updated: 2/19/2017